The peak power analyzer, a new microwave tool - includes related article on multilayer shielding to protect microvolt signals - Technical

Gallium arsenide sensor design, a new calibration approach, switched amplification and processing of the envelope signals, leveraged digital oscilloscope technology, and microprocessor control provide calibration-free, accurate pulsed microwave power measurements.

Microwave signals used in radar, telemetry, navigation, and communications applications are typically pulsed. A Doppler radar, for example, operates on the basis of pulse modulation and is critically dependent on the characteristics of the pulse power envelope, such as peak and average power, rise and fall times, pulse width, duty cycle, and pulse repetition rate. For satellite communication links in TDMA (time division multiple access) mode, the concerns are absolute and relative power levels and delays between individual pulses within a frame or from frame to frame. Even if the microwave signal is only frequency keyed or phase switched, like a QPSK signal in a digital radio application, it is often important to measure power overshoot and ringing at transition times.

The performance of systems like these can be determined to a large degree by accurate power and time measurements of the microwave pulse envelope. One stage earlier, the design, performance, and stress of the components making up such systems can be characterized by pulse parameters such as rise time, overshoot, peak power, power compression, spike leakage, pulsed gain and reflection, and gain suppression. Delay measurements are also important, not only along the microwave signal path, but also between the control signal and the microwave pulse response. Generally, the microwave response of a system or component to control stimuli is relevant in analyzing pulsed performance. Common Ways to Measure Pulsed Microwave Power What are the common ways to measure pulsed microwave power? The simplest approach is the use of a thermocouple sensor and a power meter like the HP 437B. This combination reads the average power of the microwave pulse train. Knowing the pulse duty cycle and assuming a rectangular pulse shape, the HP 437B can calculate the pulse-top power.

The spectrum analyzer can also be a tool for measuring microwave pulses. For correct amplitude determination, the desensitization factor needs to be added to the displayed amplitude and the user has to distinguish between line and pulse spectra. The spectrum analyzer yields information on pulse repetition frequency, pulse width, and pulse power. The accuracy of the latter two measurements again depends on the assumption of rectangular pulse shape. Either method is highly inaccurate if the pulses show significant overshoot, ringing, or droop, and wouldn't work at all with an irregular pulse train like the pulsed frames of TDNR signals. High-bandwidth oscilloscopes can be used in these cases. The HP 54124T, for instance, is capable of displaying the RF signal directly up to 50 GHz. While the sampling oscilloscope has plenty of speed to observe the rise time and ringing in the applications mentioned above, it falls short with regard to accuracy and dynamic range.

Diode Detector with Oscilloscope

None of the above solutions can comprehensively satisfy the measurement needs indicated at the beginning of this article. What is missing is an instrument that accurately detects and displays the power envelope of complex pulsed microwave signals, allows instantaneous power measurements on the pulses and convenient measurements of pulse parameters, all at an economic price. A potential solution has been in use on benches for a long time: a microwave detector connected to a 50-ohm oscilloscope input. However, it is difficult and cumbersome to make meaningful measurements on the displayed pulse with this setup. The video output of the detector is far from being an accurate replica of the microwave pulse envelope, voltage or power. The causes reside in part in the nonideal detector, and in part in the limitations of the oscilloscope.

Detector Nonlinearity. The diode detector has a square-law response at low power (the video voltage is proportional to the square of the RF voltage), a linear response at high power, and a wide transition range in between (approximately - 15 to + 10 dBm). Even if the user manages to calibrate the top level of the displayed signal with a substituted known RF signal, any measurement involving other levels, such as rise time, is highly inaccurate.

Detector Temperature Dependence. The temperature sensitivity of the diode detector is high and complex, changing rapidly with signal level and temperature. For instance, a 10 degrees C change from calibration can cause errors in power measurement as great as 50%. Very large errors can result from connecting the detector to a device under test that is hot or wanning up.

Frequency Response of the Video Output. The diode junction capacitance and device lead inductance are the major device parasitics affecting the frequency response of the detector output. The junction resistance also changes at high signal levels, which indirectly causes the frequency response to vary with the signal level to some degree. The unflatness of a low-barrier Schottky diode detector in X band may be around [+ or -] 0.3 dB, which translates to a [+ or -] 7% error in power.

Mismatch Error. Mismatch represents a major error source. It is proportional to the product of the reflection coefficients of the detector and the device under test (DUT). A typical low-barrier Schottky detector (SWR < 1.5) connected to a DUT with an SWR of 2.0 may give a maximum mismatch error of [+ or -] 13%. Unless the detector and DUT impedances are accurately known in magnitude and phase, this error cannot be calibrated out. Microwave Signal Harmonics. The microwave signal and harmonics of the signal are detected as a vector sum when the detector operates in the linear range. A -20-dBc harmonic may therefore cause as much as a 21% error in power measurement.

Slow Video Response. A detector requires a RF bypass capacitor on the video side (see Mg. 1). In combination with the capacitive loading of the oscilloscope, this capacitor slows down the rise time of the pulse envelope signal. The time constant can be decreased by choosing a low-impedance termination for the oscilloscope input, say 50 ohms. Unfortunately, the lower impedance also causes a corresponding decrease in detector sensitivity because the diode detector at low signal levels represents a current source.

Limited Oscilloscope Sensitivity. The low video signal levels obtained with a low-impedance termination are two orders of magnitude below typical oscilloscope sensitivities.

This list of obstacles doesn't yet include the problems of accurately calibrating the measurement. Nevertheless, the concept of envelope detection is viable and holds the promise of an economic solution. It requires only one broadband microwave component to translate the pulsed microwave signal to a video signal where it can be processed at much lower cost.

Commercial solutions based on the principle of diode detection have been on the market for some time. For example, the HP 8900C/D peak power meters use diode detectors, but represent only partial solutions. More advanced are several non-HP peak power meters which address the detector deficiencies with a calibration source and limited calibration processing.

New Peak Power Analyzer

The HP 8990A peak power analyzer (Fig. 2) is a new type of instrument that represents a comprehensive solution to the problem. The design team took a fresh look at the challenges of diode detection. Our goal was to transform the inaccurate, cumbersome bench setup into a carefree product that measures accurately and meets the complex measurement requirements of modem microwave systems. The use of GaAs IC technology in the sensor design, a new calibration approach, switched amplification and processing of the envelope signals, broad leveraging of modem digital oscilloscope technology, and extensive use of microprocessor power in signal calibration and processing accomplished this task.

Fig. 3 is a block diagram of the HP 8990A peak power analyzer.

Sensor A new approach was taken in the sensor design.1 It involves a GaAs-based diode technology, integration of a balanced circuit on GaAs, and a new calibration scheme (see the article "GaAs Technology in Sensor and Baseband Design," page 90). The use of planar doped barrier diode technology improves the frequency response and consistency of the diode characteristic.

A balanced diode circuit including the terminating load is implemented as a GaAs IC. The integration of the detector circuit minimizes mismatch loss, thanks to the very low parasitic reactances on the IC. The balanced circuit effectively minimizes the potential problem with even-order harmonics of the RF signal. Both integration and balanced design help minimize the effect of thermoelectric voltages, which could otherwise mask low-level signals.

A three-dimensional calibration scheme takes care of the problems of nonlinearity, temperature dependence, and frequency response. Extensive calibration data is taken over a wide range of power levels, frequencies, and temperatures. Condensed as a matrix of coefficient sets, the calibration data is stored on an EEPROM supplied within the sensor. When a sensor is connected to an analyzer, the analyzer reads the calibration matrix and reconstructs a precise curve of sensor output voltage versus input power for a given frequency and temperature. The frequency is entered by the user, and the temperature is measured by a thermistor chip close to the detector IC. The analyzer continuously reads the thermistor chip and automatically triggers the rebuilding of the sensor curve for any minor temperature shift. This feature of carefree calibration is especially helpful when a sensor at room temperature is connected to a hot device under test, such as a power amplifier (see the article, "Automatic Calibration for Easy and Accurate Power Measurements," page 95).

Currently, three sensor versions cover input frequencies from 500 MHz to IS GHz, 26.5 GHz, and 40 GHz, respectively. Sensors are specified from 20 dbni to -32 dBm and are usable down to - 40 dBm.

Sensor Amplifier. The problem of slow video response must be addressed right at the detector output with the sensor amplifier. The system goal of < 5-ns rise time requires special measures to deal with pulse flatness as well as drift and low-sensitivity issues. Conflicts between speed (video signals range from dc to 10-ns-wide pulses) and dc stability are resolved with a split-path design (see article, page 90). A microprocessor-controlled chopping circuit ahead of all dc-coupled circuits performs an autozeroing function at appropriate breaks in the acquisition process. This process, which is transparent to the user, allows the system to correct offsets and offset drift with temperature along the dc-coupled signal path. The dc and fast paths are reunited before leaving the sensor head. Both ends of the standard 5-foot sensor cable need to be well-matched to avoid video signal reflections.

Baseband Circuits. Moving from the sensor to the instrument, the video signal undergoes 94 dB of switchable gain. This gain resolves the problem that the limited sensitivity of an oscilloscope would present with microvolt signal levels. The dc-coupled amplifiers, switched with GaAs switch ICs, fulfill a dual task. First, they project the sensor video signal in coarse gain steps into the limited dynamic range of the track-and-hold circuit and later the analog-to-digital converter (ADC). Second, they limit broadband noise. The bandwidth of the amplifiers changes from > 150 MHz for higher signal levels to 2.5 kHz for the lowest power decade. Correspondingly, the system rise time increases from a specified < 5 ns to < 250 fts (see article, page 90).

Acquisition and Digital Signal Processing Circuits. The acquisition circuits following the baseband circuits and the digital signal processing circuits are to a large degree leveraged from the digital oscilloscope technology of the HP 54500 family. The pulse envelope signal, roughly scaled by the baseband circuits, is subsequently sampled by a track-and-hold circuit at a rate of 10 MHz (see Acquisition Process" later in this article). The hold level of the track-and-hold output then passes through a postamplifier with 15 fine gain steps. In combination with the baseband gain blocks, the postamp guarantees that any input level in the specified -32-to-+20-dBm range can fill the ADC window.

The 8-bit resolution of the flash converting ADC represents a bit range of 0 to 255. This translates to 24 dB of dynamic range referenced to the sensor input if the sensor were to operate solely in the square-law range, or to 48 dB for operation strictly in the linear range. The wide transition range between square and linear operation causes most applications to fall between these numbers. The ADC output is captured by the 2K-byte-wide circular acquisition memory.

The task of accurate time placement of the received pulse envelope is carried out by the powerful trigger and time-base ICs. A 40-MHz crystal oscillator is responsible for the time-base accuracy of 0.0050/o. A 68000 microprocessor controls the signal processing and the monochromatic 9-in display. The control code resides on a separate memory board and takes up roughly 600K bytes of ROM space.

Dual Sensor and Trigger/Oscilloscope Channels. The peak power analyzer is equipped with two sensor channels: channels 1 and 4. This facility allows pulse comparisons and delay measurements at different probing points along a microwave path or between systems. With the capability of displaying ratios of the channel inputs, the HP 8990A can measure pulsed gain and pulsed return loss (using external directional couplers). The sensor channels have internal triggering down to - 30 dBm and a trigger bandwidth of 1 MHz.

Multiplexed with each sensor channel is a video input channel. The purpose of the video channels (channels 2 and 3) is twofold: they can be inputs for external trigger signals when fast triggering (bandwidth <100 MHz) is required, and they also serve as oscilloscope inputs with 100-MHz bandwidth and limited sensitivity (100 mV/div to 500 mV/div). With these channels, the HP 8990A can simultaneously display control signals and the resulting microwave pulse envelope and measure delay times between them. It can also measure transfer characteristics like the power-versus-control-voltage sensitivity of a pulse modulator.

Sensor Check Source. A built-in source provides a pulsed or CW signal of + 10 dBm [+ or -] 0.5 dB at 1050 MHz. This serves two purposes. First, it acts as a source to verify the operation of a sensor. Peak power sensors are frequently used around high-power signals. The sensor cheek source is a convenient signal for checking the sensors if the user suspects sensor damage after an inadvertent connection to a high power level (damage level is specified at 1W peak power for 1 [micro-s], not to exceed 200 mW average). Second, the check source supplies a signal for time calibration of the trigger circuits and the timing between the four channels.

Acquisition Process

The time-base portion of the acquisition process is a duplicate of that found in the HP 54500 family of oscilloscopes, while the vertical hardware processing was modified to accommodate the greater dynamic range requirements of the IIP 8990A.

The sampling method used in the peak power analyzer is random repetitive sampling. In contrast to real-time sampling, where the sampling rate must be at least twice the highest frequency of the digitized signal (Nyquist rate), random repetitive sampling can sample at less than the Nyquist rate and still avoid aliasing.[2] Consequently, lower-speed circuits can be used in random repetitive sampling to achieve the same nominal bandwidth as real-time sampling.

The transition from a higher-bandwidth circuit requirement to a lower bandwidth is achieved in hardware through the track-and-hold diode bridge. Ideally, the bridge is modeled as an SPST switch operating at 10 MHz, closed during the track mode for 50 ns and open during the hold mode for 50 ns. In the track mode, the capacitor tracks the input signal, and in the hold mode the capacitor is isolated from the input signal. The charge residing on the capacitor during the hold time is a remnant of the input signal immediately before the bridge opened. During this 50-ns hold period, the 8-bit flash ADC digitizes the waveform. Conversion is initiated at a time when the sampled signal has settled.

The practical bandwidth limitations in the track-and-hold process are a nonzero hold capacitance and a finite diode switching time. For the HP 8990A, this translates to approximately 2-ns rise and fall times or a 175-MHz bandwidth. Random repetitive sampling provides 100-ps resolution at the fastest time-base settings. This theoretically translates to an effective sampling rate of 1/100 ps 10 GHz.2 Of course, this bandwidth is not realized because the track-and-hold circuitry limits the speed and thus acts as a surrogate anti-aliasing filter. The 100-ps resolution results from the time-base IC and the fine interpolator circuit, which time-stretches the uncertainty of one 40-MHz clock cycle. The 40-MHz clock is used to count the separation between the trigger event and the nearest data sample. Hence the possible error is from zero to one full clock cycle or equivalently from 0 to 25 ns. Instead of truncation, this residual time is fed into a time-stretcher circuit which accurately expands the time duration of the signal. The stretched signal is then counted with the same 40-MHz clock. The resultant uncertainty of one full clock cycle is divided by the stretch ratio, which is 250 in the HP 8990A. Hence the equivalent uncertainty (one clock cycle) is 1/(40 MHz)/250 = 100 ps.

The input signal is continuously sampled at 100-ns intervals for the fastest time-base settings. The sampled data is then successively placed into a circular acquisition RAM that has 2048 frames (see Mg. 4). At the end of an acquisition cycle, the RAM data's location relative to a trigger event that occurred during the acquisition cycle is determined. The samples are taken at exactly 100-ns intervals and therefore only one data sample's position relative to the trigger event need be determined to place the data appropriately in time. Both pretrigger and post-trigger data is gathered for each trigger event. The proportions of pretrigger and post-trigger data are determined by counters, which are set according to the timebase range and delay settings and the choice of left, center, or right screen placement.

The sampling signal and the input signal are asynchronous, so eventually all of the time slots, or buckets, will be filled with samples. Since the samples are randomly skewed in time between successive trigger events, no missing data or holes result in the recreated waveform. Thus a requirement of the input signal is that it be repetitive with a stable trigger event. For slower timebase settings the time buckets will be greater than 100 ps wide, reflecting the display's finite number of horizontal bits.

Continuous random repetitive sampling offers major advantages over sequential sampling. In sequential sampling only one sample per trigger is taken, with each successive trigger having an increased sample delay. In random repetitive sampling, data is continuously acquired at the sample rate, thus achieving a much faster display and providing pretrigger and post-trigger data as well. While the oscilloscope design is mainly aimed at the analysis of repetitive signals, it is also capable of capturing single-shot events. The 10-MHz sampling rate records the event with sample points every 100 ns. With a criterion of 10 sample points per event, the HP 8990A offers a single-shot bandwidth of 1 MHz.

Microwave Pulse Measurement Features

Digital signal processing makes full use of the powerful timing and trigger ICs and the properties of random repetitive sampling. The resulting features have already found wide acceptance in the HP 54500 digital oscilloscope family, whose feature set was heavily leveraged in the HP 8990A. The following are some of the capabilities that are most important for microwave pulse measurements. Time Windowing. The user of a peak power analyzer often needs to analyze a detail on a single pulse while keeping the full pulse train in view. Time windowing provides this horizontal zoom capability and allows measurements within the time window.

Trigger Conditioning. Many microwave applications present complex trigger situations that require more than a simple edge trigger function to achieve a stable display. Trigger holdoff prevents recurrent triggering on the subsequent edges of nonperiodic pulse trains, pulse packets, or bursts. Pattern trigger helps to specify a particular pulse within a frame of pulses to be triggered on, a useful feature when chasing sporadic misfires of a microwave transmitter, for example. Trigger delay, specified in time or pulse count, can be useful on long pulse trains to zoom in selectively on a particular pulse.

Persistence and Envelope Mode. Radars often operate with pulses that are extremely narrow compared to their pulse repetition interval. The duty cycle can be 0.01% or less. These signals are quite a challenge to find since most of the time the pulses fall between sample times and are not captured in the limited number of time buckets. With infinite persistence and envelope mode, they will eventually appear in a single pixel width and can then be expanded by using time windowing. Random repetitive sampling is a great advantage in this situation; a system based on sequential sampling would take a prohibitively long time to plot out such a low-repetition-rate signal. Averaging. As lower-level signals are detected, broadband noise increasingly widens the trace of the amplified signal. Choosing a narrower bandwidth, if possible, cuts down on noise, but also slows down the system rise time. Averaging in the context of random repetitive sampling means averaging sample points associated with the same point in time with respect to the trigger event, but from different acquisitions. With increasing averaging, pulses hidden in noise emerge and take shape. The digital averaging process filters noise like a low-pass filter with one important difference: it doesn't result in a rise time degradation.

Ratioing Channel inputs. The Waveform Math menu allows the user to display the ratio of any two channel inputs, in addition to performing many other useful functions. For example, the ratio of the two sensor channels can conveniently show pulse compression when probing the input and output of a limiter, or the transfer function (mW/V) of a pulse modulator can be displayed as a ratio of a sensor channel to a video channel.

Amplitude@Time Markers. The HP 8990A provides not only amplitude and time markers, but also amplitude@time markers. These denote power (or voltage for channels 2 and 3), power difference, and power ratio for a start time and a stop time. The feature is useful in determining power or gain variations along a pulse, such as pulse droop.

Thirteen Measurements. Automatic level measurements on microwave pulses, including pulse peak, average, or pulse-top power, and time measurements on pulses, including rise time, pulse width, and duty cycle are implemented as simple blue-key shift functions. The reference levels for these measurements, that is, pulse top (100%) and pulse base (0%), are histogram-based according to IEEE standards.

Applications

The combination of two microwave sensors and two video inputs, able to make complex microwave and video pulse-measurements in one instrument and relate them in level and time, promises broad applications in many areas.

Radar Components and Systems. Starting out with a traditional area of peak power measurements, the characterization of the power transmitter is central to radar performance. Peak and average power, rise time, overshoot, and droop are standard measurements at the output of the transmitter. With the use of couplers, the dual microwave channels facilitate measurements of pulsed gain, gain compression, and pulsed return loss. Coupler losses can be compensated numerically in the HP 8990A.

On the receiver side, the receiver protection limiter needs to be characterized in terms of spike leakage and spike compression. Fig. 5 displays the output of a limiter. The time windowing feature is used to focus on the spike detail, and markers spell out the spike leakage in dB. The ratio of the input of the limiter to its output would show the power compression along the pulse. At the system level, the peak power analyzer performs delay measurements between, for instance, pulse drive and transmitter output. The concurrent and time-calibrated display of the drive signal and the microwave pulse avoids cumbersome calibration tasks.

Analyzing the transfer characteristic of the pulse modulator is another example of the combined use of sensor and video inputs. The transfer characteristic can be displayed as a ratio of the microwave channel to the video channel and examined for linearity.

Complex Communication Signals. The powerful trigger capability of the HP 8990A comes into play when signals like the pulse bursts of a TDMA (time division multiple access) system are to be evaluated. For instance, trigger holdoff stabilizes the display by inhibiting recurrent triggering on subsequent edges. Trigger delay allows the user to select, say, the 231st pulse of a long pulse train. Trigger pattern lets the user trigger on specific pulses, like glitches, within a train or burst. The dual time base is useful for observing and measuring time or power from burst to burst and simultaneously on individual pulses within a burst (see Fig. 6).

Not so obvious is the application of the peak power analyzer to communication signals keyed in frequency or phase. The peak power analyzer can be used here to measure accurately the power glitches and changes that accompany phase and frequency switching. More complex digital modulation formats, like the 16QAM signal of a digital radio, can be examined with regard to level transitions, overshoot, and power (amplitude) compression. Infinite persistence is useful for recording multiple traces (see Mg. 7).

Transient Response of Components. Transient measurements on microwave components usually require cumbersome calibration to account for control signal and instrumentation delays. The time calibration provided in the HP 8990A between the sensor channels and the video input channels and the simultaneous display of both the control stimulus and the microwave response make this task easy.

The pin switch in Mg. 8 is turned on by the falling edge of the control signal. A single blue-key operation measures the delay between the mesial points of the falling control signal (lower display) and the rising microwave pulse at the pin switch output. The wide pretrigger range makes it possible to view the full turn-on region of the transition.

Conclusion

The applications listed above give an indication of the versatility of the HP 8990A peak power analyzer. The integration of advanced sensor and calibration technology with the trigger and signal processing power of HP's advanced digital oscilloscopes results in an instrument that delivers carefree and accurate measurements on microwave pulses and driving control signals.

Acknowledgments

We would like to thank the HP Colorado Springs Division's HP 54500 team for their time and technical support. The enthusiastic collaboration of Helen Muterspaugh and her firmware team, and on the hardware side, Laura Whiteside and Yvonne Utzig, was key to the successful leveraging of digital oscilloscope technology and gave a powerful initial boost to the peak power analyzer development. Acknowledgments to the production engineering team, headed by Jay Anderson. Their efforts were critical for the fast-paced introduction of the project. Credit also goes to Darren McCarthy who was the driving force in reliability testing, to Allen Edwards for his technical contributions and his support as section manager, and to Jim Jensen who led the firmware effort.

References

1. D. Scherer, "Designing Sensors to Read Peak Power of Pulsed Waveforms," Microwaves & RF, February 1990.

2. R.A. Witte, "Understanding Digitizing Oscilloscopes," RF Expo Digest, 1989.

COPYRIGHT 1992 Hewlett Packard Company
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